The present invention relates to the technique field concerning professional telecommunication systems, and more particularly a broad band transmitter for a signal consisting of a plurality of digitally modulated carriers.
The use of the radio frequency spectrum in telecommunications is governed by international standards which assign specific frequency bands to given services, both public and private. Inside these bands, services are generally organized in order to take advantage of the band occupation to the best extent, for instance, dividing the same into a plurality of contiguous channels. We have number of examples on this matter. A first example is represented by telephone radio links, where thousands of telephonic channels are multiplexed among them, in frequency or in time, and the multiplexed signals are used to modulate the relevant carriers of a same number of radio channels, arranged in order to result contiguous within a microwave band. A second example is given by the Pan-European communication system, hereinafter defined with the acronym GSM (Global System Mobile), based on the time share use of as much as 124 carriers, 200 KHz spaced among them, digitally modulated according to a GMSK scheme (Gaussian Minimum Shift Keying), and individually transmitted within a 35 MHz band (Extended GSM) placed around 900 MHz. The reference to the GSM system is desired since, being the same an essentially digital system, it is the field of preferential application of the transmitter according to the subject invention. As it is already known, by digital modulation we mean a modulation scheme where the parameter, or the parameters, characterising the modulated carriers assume only a discrete number of values; in the GSM, like in the most advanced telecommunication systems, the carriers are orthogonally phase modulated starting from a modulating signal consisting of bursts of information bits.
In any type of transmitter for digital signals, in addition to the usual filtering of the image band generated by the radio frequency converter and of the residual of local oscillator, it is necessary first to filter the replicas of the base band spectrum caused by the conversion of the digital signal to the analogue form, the sole possible for radio transmission. FIGS. 1 and 2 show what described above. In particular, in FIG. 1 we can notice that the sampling frequency fs is higher than the double of the useful band BW of the signal to sample, as defined by the Nyquist criterion to avoid spectral superimposition in the sampled signal.
In the case a multicarrier transmitter is implemented, according to the architecture that can be assumed, the above filtering can result more or less expensive. In fact, if one wants to construct a multicarrier signal of the digital type, it should be useful to sum up in a digital way the largest possible number of modulated carriers in order to avail of the speed allowed by the digital section performing such construction to the maximum extent, compatibly with the maximum operation speed of the digital-to-analogue converter. However, this operation method would involve a considerable shortening of the distances existing between the lower edge of the base band and the continuous, on one side, and the upper edge and the fs/2 frequency, on the other side. The above mentioned distances are indicated with xcex94F in FIG. 3 and have the following expression in case of simmetric allocation of BW in the Nyquist band:                               Δ          ⁢                      xe2x80x83                    ⁢          F                =                                            (                                                                    f                    s                                    2                                -                BW                            )                        2                    .                                    (        1        )            
The approaching of the useful spectrum to the continuous would complicate the radio frequency filtering to eliminate the residue of local oscillator and the image band (see FIG. 2), while the approaching to fs/2 would complicate the reconstruction filtering for the elimination of undesired spectral replicas (see FIG. 1). There is therefore a compromise between the choice of the sampling frequency fs and the bandwidth of the multicarrier signal in the first Nyquist area. Concerning the sampling frequency, it corresponds to that of a clock signal used by the digital section. Said frequency shall necessarily be higher than that resulting from the choice of two samples to represent the modulated numeric phase carrier placed at the upper edge of the broad band spectrum, since it is necessary to maintain said filtration margins. The maximum value of the sampling frequency should be at present 40 MHz approximately, limit imposed by the technology of the marketable components, while concerning the maximum band width of the useful signal, this would depend on the margin one wants to leave to simplify the above mentioned filtering. At 40 MHz frequency no limit would be imposed by the digital/analogue converter, which can easily reach a speed more than double.
It is now assumed the project of a broad band transmitter for a digital multicarrier signal, to the purpose of highlighting the difficulties encountered in a similar implementation, difficulties that up to now have discouraged this type of realization approach. In the postulated transmitter we assume:
sampling frequency 34.6 MHz;
number of channels 16;
a spacing between channels 600 kHz, corresponding to a GSM cluster size equal to 3.
With these assumptions it results that the band of the useful signal BW occupies 10 MHz approx., to be allocated in a first Nyquist area, 17.3 MHz wide. Considering to position the intermediate frequency IF at the centre of the first Nyquist area, that is: IF=8.65 MHz, we obtain that the distances xcex94F between the edges of the useful spectrum and the edges of the first Nyquist area have a value of 3.65 MHz; the margins destined to filtering are therefore very narrow.
FIG. 4 shows the GSM 11.21 specifications relevant to the emission of spurious signals (for systems operating in the GSM band). They foresee that each spurious signal emitted by the transmitter lays under xe2x88x9236 dBm in the whole frequency spectrum up to 1 GHz, except for the reception band, where it is necessary to observe xe2x88x9298 dBm. For frequencies higher than one GHz the specifications impose to emit no more than xe2x88x9230 dBm, except for the bands destined to the 1800 MHz DCS service (Digital Cellular System).
Assuming to employ a local oscillator power Pol equal to 10 dBm, to have an isolation between the local oscillator and the radio frequency in the balanced mixer that generates the frequency convertion of ISOxe2x80x94olxe2x80x94rf=30 dB, and that the gain of the whole transmission chain Gtot is 50 dB, we obtain that at the output, without filtering, the residue of local oscillator Resol is equal to:
Resol=Polxe2x88x92Isool+Gtot=10xe2x88x9230+50=30 dBm.xe2x80x83xe2x80x83(2)
In the case the residue of local oscillator falls in transmission band, it is necessary to increase the 30 dBm to xe2x88x9236 dBm, that is, a radio frequency band pass filter must be employed, which at a distance xcex94F=3.65 MHz from the edges of the band attenuates 66 dB. To obtain this, it is necessary to use two identical Chebyshev filters with 6 resonators; a similar filtering results very expensive.
In addition to the disadvantage of an expensive radio frequency filtering, the use of a low IF could involve a second disadvantage represented by the fact that conversion products generated by the non linearity of the mixer could fall in the useful band of the signal. The mixer in fact, besides generating undesired signals at the frequencies:
fOLxc2x1fcanxe2x80x83xe2x80x83(3)
produces spurious signals at frequencies:
Nxc2x7fOLxc2x1Mxc2x7fcanxe2x80x83xe2x80x83(4)
for all the combinations of M and N integer positive and negative. The width of the spurious signals decrease as M and N increase. Out of these spurious products, those, which can highly disturb the radio frequency signal, are those of lower rank, because they have higher width and, as we will see now, they can fall in the useful band.
Let""s consider the case N=1 and M=2, that is
f1MD1.2=fOLxc2x12fcanxe2x80x83xe2x80x83(5)
FIG. 5 shows the radio frequency spectrum situation for the upper side band only, in case of spectrum consisting of only two frequency channels fcan1 and fcan2, one at the beginning and the other one at the end of the useful spectrum BW:
BW=fcan2xe2x88x92fcan1xe2x80x83xe2x80x83(6)
It can be easily checked that if the following condition is true:
fcan1 less than BWxe2x80x83xe2x80x83(7)
the product due to the non-linearity of the second rank (IMD2) of the mixer falls in the band of the signal to transmit. From the measurements made on some samples we notice that even using double balanced mixers we always have a second order contribution of xe2x88x9250 dBc at least, which when amplified does no more fall under the specifications of FIG. 4 and in any case does not comply with the specifications concerning intermodulations in transmission band, therefore it is necessary to avoid that said contribution can fall in the band of the signal to be transmitted.
In the 16-carrier transmitter of the case considered, the useful spectrum occupies the frequency band between 3.65 MHz and 13.65 MHz, and BW=10 MHz, from which:
xe2x80x83fcan1(3.65 MHz) less than BW(10 MHz),
and the condition (7) results therefore checked, consequently the spurious products irremediably fall in the band to be transmitted. A method to avoid that this happens could be that to employ a higher intermediate frequency IF, that can be obtained with a higher clock frequency for the digital section constructing the multicarrier signal. Operating in this way, at equal BW band, we can obtain a higher distance xcex94F of the useful spectrum BW from the continuous, until the condition (7) is no more checked, but the contrary condition results inspected:
xcex94F=fcan1 greater than 10 MHz.xe2x80x83xe2x80x83(8)
Unfortunately, the present technological limits of the components employed do not allow such a solution.
The disadvantages described above have until now discouraged the implementation of a digital transmitter of the postulated type. In fact, in the BTS (Base Station Transceiver) of the major manufacturers of mobile systems, where such a transmitter could be profitably employed, what is actually used is a multicarrier transmitter consisting of a plurality of independent mono-channel transmitters, coupled to a unique or to a limited number of antenna. In this way, since a digital section of the type of the postulated transmitter is not present, also the relevant clock problems disappear, as well as those concerning the filtering of the analogue-converted signal replicas, being the transmitters of the narrow band type. It is also used a second intermediate frequency to further simplify the radio frequency filtering.
FIG. 6 shows a multicarrier transmitter actually employed in a base transceiver station, or BTS, of a mobile communication system GSM. For convenience, only two out of the N identical mono channel equipped transmitters are shown, identified RFTX1 and RFTXN, respectively. The input of the transmitters RFTX1, . . . , RFTXN is reached by relevant bit strings at 270 Kbit/s that convey the transmission burst relating to a same number of communication channels CH1, . . . , CHN assigned to the users in conversation. These signals inside the relevant transmitter reach a modulator MOD operating the GMSK modulation of a sinusoidal carrier at 200 KHz in digital form, giving at output a digital signal reaching the input of a digital/analogue converter DAC. The sampler of the DAC is controlled by a CK signal having a value such for which the signal spectrum (PAM) coming out from the DAC has side bands around a first intermediate frequency IF1. The converted signal is filtered by a first band pass reconstruction filter FPB1 that selects the desired band 200 KHz wide. The signal coming out from FPB1 is then sent to an input of a first balanced mixer MIX, reached also by a first local oscillator signal OL11 at a second intermediate frequency. The signal at second intermediate frequency coming out from MIX1 is filtered again by a second band pass filter FPB2, which selects the desired side band. The signal coming out from FPB2 is then sent to an input of s second balanced mixer MIX2 reached also by a second local oscillator signal, OL21, . . . , OL2N respectively, relating to transmitters 1, . . . , N, for the radio frequency conversion. The signals OL21, . . . , OL2N differ among them in frequency and are generated by a same number of PLL. In each transmitter the signal coming out from MIX2 is filtered by a relevant image filter FIM1 that eliminates the image band and the local oscillator residue from the radio frequency spectrum, and then sent to a radio frequency power amplifier LPA operating in class A. Amplified signals are newly filtered by relevant channel filters FCH1, . . . , FCHN, having high Q, at whose output a same number of radio frequency signals are present RF1, . . . , RFN, that shall be transmitted. In cascade to the N transmitters, a block RFCOMB at N inputs is placed for the signals RF1, . . . , RFN, that couples the signals present at its inputs to a reduced number of directive antennas ANT1, ANT2 and ANT3, assuming a xe2x80x9ccorner excitedxe2x80x9d configuration, obtaining by this a transmission multicarrier signal of the TDMA type.
The chain of blocks forming the RFTX1, . . . , RFTXN transmitters is subdivided into contiguous sections identified with BB, IF1, IF2 and RF to distinguish the operation, in base band, at the first intermediate frequency IF1, at the second intermediate frequency IF2, and at radio frequency RF, respectively.
FIG. 7 shows the operations accomplished on the frequency spectrum by the blocks belonging to sections BB, IF1, IF2 and RF of the transmitters of FIG. 6. In (A) it is shown the band of the signal at intermediate frequency IF1 coming out from the filter FPB1, without the replicas generated by the DAC converter; in (B) the band of the signal shifted to IF2 coming out from the filter FPB2; and finally in (C) the RF signals coming out from the image filters FIM1, . . . , FIMN, without the image band. The correct positioning of channels CH1, . . . , CHN in the spectrum of the GSM band is due to the different shift in frequency made by the mixers MIX2, controlled by different frequencies for different channels, together with the combination to RF made by the multiple coupler RFCOMB.
As it can be noticed in (C), thanks to the double frequency conversion, both the residue of local oscillator, and the undesired side band are distant from the signal to transmit and therefore more easy to filter.
The architecture of the transmitters of FIG. 6 corresponds, more or less, to that adopted in the BTS of the major manufacturers of mobile systems, this does not mean that it is without inconveniences. In fact, we can immediately notice the great complexity of the whole, essentially due to the mono-channel architecture, which requires a repetition of all the blocks of the transmitter for each group of the eight channels time-share assigned to a single carrier. Consequently, the same applies to the structure of the radio frequency combiner RFCOMB, which shall have an input port for each carrier used. In addition to the above, there are also the necessary redundancies, which become increasingly expensive as the circuitry complexity increases. Furthermore, due to the conversion to the second intermediate frequency, a mixer is required for each single transmitter, a PLL and a band pass filter in excess, compared to the use of a single intermediate frequency. A similar architecture results too expensive and cumbersome, since the analogue portion is prevailing.
EP 0 534 255 discloses a transceiver system capable of simultaneously servicing multiple digital channels and it includes:
means for generating digital carrier signals;
means for modulating said digital carriers;
means for accumulating the modulated signals;
digital to analog converter (DAC) means common to each of the modulated signals;
means for, altering the frequency of the composite signal to a desired radio frequency band.
According to the above, the equalization process is accomplished in digital form by means of a FIR filter and this constrains the digital part of the system to work at a higher frequency than the DAC.
Therefore the object of the present invention is to overcome the above mentioned drawbacks and to indicate a process for the implementation of a broad band transmitter for a signal consisting of a plurality of not necessarily equispaced digitally modulated carriers.
To reach said objects the present invention discloses a process for the implementation of a multicarrier digital transmitter, comprising the steps of:
a) digital modulation of one, or more, parameters of single numeric carriers belonging to said multicarrier signal, obtained using data information conveyed by N sequences of transmission bits coming from a predetermined number of communication channels;
b) sum of numeric samples of said numeric carriers modulated as explained in the step a), obtaining a sequence of samples at sampling frequency fs of said multicarrier signal;
c) conversion to the analogue form of the above mentioned sample sequence, said process being characterized by the following additional steps:
d) equalization through an analogue filter of the       sin    ⁢          (              π        ⁢                  xe2x80x83                ⁢        fT            )            π    ⁢          xe2x80x83        ⁢    fT  
xe2x80x83function that envelops the frequency spectrum of said sequence of analogue samples at the output of the digital to analog converter, being T the clock period of the digital to analog converter and f the independent variable frequency;
e) band pass filtering at intermediate frequency to select an n-th replica of the base band spectrum of said multicarrier signal;
f) radio frequency conversion of said replica selected at intermediate frequency;
g) radio frequency band pass filtering for the selection of a side band of the radio frequency converted signal, to amplify and couple to at least one transmitting antenna.
Another object of the invention is a transmitter implemented according to the above process, as disclosed in claim 10.
According to the above, the gist of the present Invention is in defining a system including:
means for digital modulation of a multicarrier signal;
means for summing the numeric samples of the modulated carriers;
means for the conversion to the analogue form (DAC) of the signal;
means for the equalization, through an analog filter, of the sin(xcfx80fT)/(xcfx80fT) function that envelopes the spectrum of the analogue signal
means for band pass filtering at intermediate frequency to select an n-th replica of the base band spectrum of the multicarrier signal;
radio frequency conversion and radio frequency filtering means for the selection of a side band of the radio frequency converted signal.
In the claimed system the analogue filter permits to the digital part of the system to work at lower frequency than the DAC, in addition an analogue filter is easier to implement, more reliable and cheaper than a FIR filter.
In the claimed process and system, the major advantage of the frequency shift due of the selection of an n-th replica, is a shift to an intermediate frequency, the signal is then converted to a radio frequency by a mixer.
Another benefit introduced by the analogue equalization is that it can be accomplished through the resetting (TG) of the sampled converted to analogue, this permits to choose the duration of the rectangular sample in a sample interval. So doing it is possible to choose where to put the null of the sin(xcfx80fT)/(xcfx80fT) function that envelopes the spectrum of the analogue signal in respect of the desired replica of the signal. Therefore it is possible to choose what course of the sin(xcfx80fT)/(xcfx80fT) function interests the desired replica. This is no possible to accomplish in digital form because it is possible to put only a finite number of zero samples between two samples of the digital signal, so the null of the sin(xcfx80fT)/(xcfx80fT) function is always placed corresponding to integer multiples of the sampling frequency.
Therefore, the substantial difference between the systems disclosed in EP 534 255 and the claimed process and system is the analogue equalization of the claimed system that is capable to recuperate the sin(xcfx80fT)/(xcfx80fT) function that envelopes the spectrum of the analogue signal.
The great advantage of such a transmitter lies in its simple architecture, completely assigning to a digital section the construction of a multicarrier digital signal, of the TDMA type, obtained summing up different digitally modulated carriers. Said section employs to the purpose a clock signal whose frequency is necessarily higher than that resulting from the choice of two samples to represent the modulated numeric phase carrier placed at the upper end of the broad band spectrum, since it is necessary to maintain the above mentioned filtering margins.
The invention solves all the technical problems highlighted, inevitably involved in a similar architecture, including radio frequency filtering ones, which would otherwise arise from the lack of a second intermediate frequency. To this purpose, it is converted at radio frequency an n-th spectral replica of the multicarrier signal converted to analogue, preferably the second one. This enables a more unconstrained radio frequency filtering and enables to avoid that spurious conversion products of the upper rank fall into the useful band of the transmission signal (thing that would render practically impossible the realization of RF filters). Since the spectrum of the signal coming out from the DAC is enveloped by a function of the sin(xcfx80fT)/(xcfx80fT) type, the choice of an n-th spectral replica involves the fact to introduce a width equalization in the course of the same.
The equalization can be made for instance after the conversion to analogue through a block having transfer function with course of the (xcfx80fT)/sin(xcfx80fT) type, in the frequency area where one wants to drawn the signal followed by an amplifier block that has the purpose to recover the attenuation introduced by the previous equalizer block.
With continued reference to the invention, the equalization can be alternatively made through a simple interpolation (zero insertion) of the multicarrier digital signal before the D/A conversion or after the conversion to analogue.
In this context, the periodical repetitions of the base band spectrum are defined xe2x80x9creplicasxe2x80x9d: the base band spectrum is therefore indicated by the term xe2x80x9cfirst replicaxe2x80x9d, its repetition included between fs/2 and fs is identified xe2x80x9csecond replicaxe2x80x9d, and so on. The n-th replicas characterized by even n have the peculiarity to be specular versus the base band spectrum (see FIG. 1). If, as it occurs in a preferred use, the replica to be converted at radio frequency is the second one, it is necessary to select, at radio frequency, the lower side band towards the local oscillator frequency, since in this way the second inversion of the spectrum operated by the mixer is used to restore the same in the condition of the base band spectrum. Advantage is taken also from the fact that, thus making, the non desired side band, generated by the mixer, never falls in the spectrum destined to the GSM reception, where the specifications are more stringent (see FIG. 4).
A brief description follows of the zero insertion equalization before the conversion to analogue and its variant to analogue.
The zero insertion technique has the purpose to equalize the signal spectrum, reducing to Txe2x80x2 the duration of the samples coming out from the DAC with sampling period T=1/fs through an interpolation made on the same (the relation Txe2x80x2=T/n with n=positive integer number applies), at the DAC input or output without distinction. This enables to place the zeroes of the sin(xcfx80fTxe2x80x2)/(xcfx80fTxe2x80x2) function that envelopes the frequency spectrum at the output of the DAC, at the frequencies 2 nfs=1/Txe2x80x2.
The architecture of the transmitter according to the subject invention is such that the blocks of the transmitter placed downstream the D/A converter are unique for the multichannel complex, contrarily to what occurred in the transmitter of FIG. 6. Also, said blocks do not include a second mixer for the conversion to a second intermediate frequency, with the respective PLL and band pass filter. Great advantages can therefore be obtained from the use of such a transmitter in a base transceiver station (BTS) of a mobile GSM, or DCS system.
Due to what said above, we can decidedly assert that the architecture of the multicarrier transmitter according to the present invention is able to considerably reduce the production costs of the BTS and to improve the repetition capability and reliability, since the possibilities offered by the less expensive digital techniques concerning signal processing are exploited to the maximum extent, saving the analogue as much as possible.